AUDIOPI YANG KUNCI QUEST UNTUK SISTEM RUMAH ULTIMATE Apakah Anda merindukan masa lalu - apakah Anda dengan senang hati mengingat suara jambul jock quartbanger Mengetuk jam dapur art-deco yang mengerikan dari dinding dengan rekaman Mercurys Antal Dorati pada tahun 1812 dan membangkitkan ooh dan ahhs Dari teman-teman bodoh yang tidak bisa percaya telinga mereka saat mendengar 35 watt besar per saluran dan suara Bob Prescotts quotCartoons di Stereoquot Pada tahun 1961, orang-orang dari kita yang dengan berat hati menentang istri atau orang tua kita dan menghabiskan 355.80 ditambah biaya 10-12 keterlaluan Kayu lapis kayu lapis kelas tinggi untuk membangun 14 kabinet kaki kubik kita sendiri, hidup dalam kebahagiaan dengan kepercayaan yang dipuji, bahwa sepasang peluru D130 dan 075 sama bagusnya dengan sistem speaker yang perlu dilakukan, rekaman musik itu tidak akan pernah bisa menantang seperti itu. Sistem, dan suatu hari nanti jika kita mendapat pengembalian pajak yang besar, kita mungkin berpikir untuk menambahkan sepasang 175DLH untuk membuat sistem tertinggi. Kami adalah elit audio - cognoscente yang memegang pengadilan bagi mereka yang mengira kami jenius karena kami dapat menyatukan Mac 60 dan preamp dan benar-benar mengatur pemerataan cakram yang benar untuk salah satu dari banyak disket rekaman perusahaan rekaman yang memotong EQ yang digunakan. Saat itu - sampai kecewa pecinta musik non-insinyur. Jika Anda menyukai saya, anak berusia lima puluhan, kemungkinan ingatan Anda akan sistem efisiensi tinggi yang pertama menggerogoti Anda dan membuat Anda bertanya-tanya apa yang ada di dunia ini tentang ribut tentang sistem speaker ready-token. Ya, JBL sudah siap digital 45 tahun sebelum digital sudah siap. Tentu saja, Anda masih bisa mendapatkan E130 dan 2402 (model model komponen lama saat ini) dan membodohi diri sendiri untuk memikirkan hal itu hi-fi, tapi jika Anda masih Tweak audio Anda pada tahun 1961, hasil dari pemikiran usang ini akan terbukti tidak menarik - suara yang Anda ingat tidak akan cukup baik lagi. Suara yang Anda dapatkan tidak sesuai dengan kenangan akan hal itu mengingat apa yang mungkin Anda dengar selama 45 tahun terakhir. Penggemar wallbanger yang baik, sejak tahun 1961, beberapa kemajuan telah dicapai dalam memahami pengalaman mendengarkan, alasan mengapa sound system tidak pernah terdengar seperti pertunjukan live, dan bagaimana memperbaiki situasi misterius itu. Kita tahu bagaimana semua perangkat keras bekerja sekarang, dan kita tahu lebih banyak tentang mengapa ada banyak cara untuk membuat sistem speaker yang buruk. Seperti yang mungkin dikatakan oleh filsuf Denmark, kuotaudio seperti filsafat di setiap langkah yang meluncur dari kulit tua yang membuat gantungan baju yang tidak berguna-onquot. Jika Anda ingin tetap berada di ujung tombak teknologi favorit Anda, Anda harus memiliki pandangan eklektik mengenai perbaikan desain masa lalu. Sudahlah kenyataan bahwa todaysquot perekaman insinyurquot memiliki rambut hijau dan tidak dapat membaca musik dan bahwa sebagian besar dari apa yang masuk ke gigitan pop cd berasal dari kotak diprogram. Agar adil, ada banyak CD yang terekam dengan baik yang dapat Anda dengarkan, dan setiap alasan untuk mengharapkan bahwa materi program yang bagus akan dibuat oleh mereka yang peduli akan musik dan kualitas audio lebih banyak daripada pemasaran massal. Meskipun JBL penjualan komponen mentah ke pasar rumahan terus berkembang sejak tahun lima puluhan, JBL sebagai perusahaan, telah melakukannya dengan sangat baik dalam audio profesional (sistem suara yang terpasang secara permanen, sistem suara tur, suara bioskop dan alat musik speaker dan Komponen) selama beberapa dekade terakhir, dan pemasar hif telah membanjiri masyarakat dengan begitu banyak pilihan pembicara siap pakai, bahwa proporsi JBL total penjualan komponen mentah ke pasar hif telah dibayangi pada titik di mana mendukung pasar tersebut. Segmen sekarang tidak menguntungkan. JBL mencintai penggemar setia, tapi waktu yang dibutuhkan untuk menjawab ribuan pertanyaan dari mereka mengimbangi penjualan yang dihasilkan. Sebagai hasil dari ini dan kenyataan bahwa hampir semua panggilan konsumen yang diterima oleh JBL Professional sekarang menjadi pertanyaan untuk membangun sistem tertinggi (walaupun kami berusaha untuk mengirim Anda ke Harman America dan menjual sistem pengeras suara konsumen 250 Ti yang spektakuler) . Saya telah memutuskan untuk menjawab semua pertanyaan Anda secara tertulis dengan harapan Anda tidak akan memanggil dan mengomel saya. Ini adalah apa yang saya pikir secara pribadi akan saya lakukan jika saya memiliki banyak uang untuk dibelanjakan di sistem rumah saya dan tidak dapat menghadapi sesekali dokter dengan sistem KrellRowlandCello yang sangat mahal, menantang hak membual. BAGAIMANA BAIK DAPAT DIGUNAKAN DENGAN KOMPONEN PRO Sejumlah besar audiophiles yang tidak puas, orang-orang aneh dan banyak orang tua JBL telah menelepon dan menulis dengan bersikeras pada rekomendasi saya untuk sistem pemutaran stereo rumah yang lebih besar yang mungkin memberikan semua kenyataan yang mengganjal perut. Meringkuk dalam posisi janin di dalam drum kick rock-n-roll. Meskipun manfaat penghancuran pendengaran yang dilakukan sendiri berhasil menyelamatkan diri, saya menawarkan apa yang saya anggap sebagai alternatif yang berguna (untuk mereka yang cenderung) untuk menyewa live band dan perusahaan tur suara ketika dorongan untuk penyiksaan diri pendengaran muncul. Sistem quotdream yang dijelaskan di sini tidak akan mengupas cat dari dinding Anda atau cukup sebagai sistem PA untuk ruangan yang lebih besar daripada bangunan perakitan vertikal biasa, namun harus memuaskan hasrat pendengaran dari punkers yang sangat berubah, disko-droid, rapper dan yang paling masokis. Penggemar metal-rock, sambil tetap menyediakan kehalusan yang adekuat untuk musik ruang barok yang halus, Hogwood Brandenburg tahunan Anda, dan rekaman efek suara serangga populer di mana-mana. KONSEKSI SISTEM DARI BERIKUT: 4 2245H 18wm subwoofer driver 2 2220H Driver midcass 15quot 2 2123H Driver midrange 10quot 2 Driver penguat 24quot 2445J 2 2382 Tanduk Bi-Radial Datar Depan 2 2405 Balancing tweeter 1 6290 power amplifier 4 6260 amplifier daya 2 6230 Amplifier daya 2 525 crossovers aktif 2 3105 crossover pasif Biaya total sistem untuk komponen ini saja, sekitar 18.000 sehingga debu dari kertas hipotek lama dan gaskan Rolls untuk perjalanan ke bank sebentar di rumah Anda. Penguat amplifier yang terdaftar akan, sesuai permintaan, menghasilkan 1200 watt ke empat pengisi daya 18quot, 1200 watt ke dua driver mid-bass 15quot, 1200 watt ke dua driver midrange 10quot dan 600 watt ke dua driver kompresi dan unit tweeter. Total on-demand power adalah 4200 watt bersih. Ini mungkin juga menuntut Anda - dari tetangga dan polisi setempat. Sekarang sebelum Anda terkesiap dan ekspektasi quotyech. Tanduk quot sadar bahwa segala sesuatu yang Anda dengar adalah sejarah dan sebagian besar salah. Model 2382 adalah tenggorokan dua inci, perangkat tipe waveguide 120 derajat dengan tingkat suar yang cepat dan suara quothorn yang hampir tidak ada yang disebabkan oleh kerongkongan non linier yang terkait dengan tenggorokan satu inci yang lebih kecil dan tingkat flare eksponensial dan wont Digunakan dalam sistem ini untuk mereproduksi frekuensi yang cukup rendah sehingga menjengkelkan pula. Ingatlah salinan katalog JBL dari tahun enam puluhan: muka gelombang kedap suara dari sudut ledakan diperkuat oleh magnet kuat dan diafragma kuadran 4quot. Tetap berpikiran terbuka jika Anda berharap mendapat ganjaran dengan tingkat tekanan suara tinggi. Salah satu kodratnya yang tidak berubah, Anda harus membuat setidaknya beberapa konsesi untuk mendapatkan manfaat tertentu (seperti tingkat suara yang sangat tinggi). Maaf, tapi Anda tidak bisa mengubah hukum fisika dengan uang. Akustik setinggi 7 kaki Anda akan terdengar seperti radio transistor 4 di sebelah sistem Anda, jadi berhentilah menggigit kuku Anda dan tulislah ceknya. Anda perlu membangun atau memperoleh (jangan hubungi kami, kami tidak dapat membantu) lemari yang akan menyediakan volume internal 20 kaki kubik untuk masing-masing pasangan driver bass 18quot, kandang terpisah 1,5 kaki kubik untuk setiap driver midbass 15quot, sebuah Sub-enclosure atau kandang terpisah dari 0,3 kaki kubik untuk driver kelas menengah 10quot dan permukaan pemasangan untuk tanduk dan tweeter. Seluruh perselingkuhan (satu anggota kiri atau kanan pasangan) mungkin berukuran antara 48 dan 60 inci, lebarnya sekitar 5 kaki, lebarnya sekitar 3 kaki dan beratnya akan banyak. Bangun kandang rendah bass dari sesuatu yang kaku seperti lempeng beton 6 inci yang diputar di sekitar cincin pemasangan woofer yang terbuat dari kayu lapis birch 14 lapis Finlandia, atau cukup gunakan kayu lapis dan penahan dua per empat yang dilem dan diikatkan di mana saja di mana saja. Anda dapat mendeteksi resonansi panel saat menabrak panel dengan palu pembentuk 2 pon Anda. Tujuannya di sini adalah membuat lemari jadi setebal beton atau setidaknya setepat mungkin. Ingatlah bahwa sistem akan terdengar lebih baik jika Anda membangun semuanya ke dalam sofits yang mengeras di dinding, jadi sebaiknya Anda memiliki sewa lama atau memiliki rumah yang ingin Anda ubah. Dimensi kotak interior yang tepat untuk selungkup subwoofer adalah 41 X 33,5 X 29 inci. Sisi 29 X 41 digunakan untuk memasang woofer. Ventilasi ducted terdiri dari dua papan, 9,25 X 29 inci dipasang di tengah antara dua woofer. Terowongan lubang slotted ini berfungsi baik untuk menyetel selungkup dan menyangga panel samping. Area terbuka ventilasi dan terowongan adalah 4,5 X 29 inci (lebar kotak), dengan kedalaman total 10 inci. Line kotak interior di semua sisi dengan satu lapisan 1 inci tebal, serat kepadatan setengah pon untuk redaman refleksi internal. Tidak ada manfaatnya, dan faktanya, mungkin ada kemerosotan dalam performa jika banyak serat kaca digunakan. Fiberglass menambahkan volume virtual ke selungkup. Kenakan masker dan sarung tangan saat Anda memasukkan barang-barang di sekitar penyangga (kecuali jika Anda memasang bracing di bagian luar kotak) atau ke interior panel lalu ambil shower dingin saat Anda selesai. Mungkin Anda tidak akan gatal dan batuk selama seminggu. Sebuah kata peringatan bagi pecinta hewan: jika Anda memiliki seekor kucing, Anda harus menggunakan layar kawat ayam di bagian dalam duktus di sela woofer untuk mencegah kucing yang ingin tahu kehilangan salah satu dari sembilan nyawa mereka saat meriam dari tahun 1812 Pembukaan membangunkannya dari tidur mereka yang nyaman di dalam kotak. Untuk pengemudi midbass, Anda harus membuat kandang yang sangat padat dan bebas resonansi untuk dipasang di atas kandang bass. Setelah memerah ke dinding, tidak masalah jika kotak Anda tidak sesuai dengan dimensi lebar dan kedalaman. Dimensi interior yang tepat dari enclosure midbass adalah 18,9 X 15,4 X 13,4 inci dan Anda memerlukan ventilasi yang terdiri dari slot 2 inci X 5 inci, memotong material 34 inci dari baffle, di suatu tempat di dekat tepi driver midbass. . Seperti halnya kandang woofer, oleskan lapisan fiberglass ke dinding interior kotak. Pengemudi midrange ditempatkan di kandang tertutup terpisah yang dimensi dalamnya 10,7 X 8,7 X 7,6 inci. Selungkup ini juga harus dilapisi padding fiberglass yang sama, dengan lapisan ekstra di bagian belakang kotak. Cara terbaik adalah membangun kandang ke perancah datar besar untuk mengakomodasi pemasangan pengemudi karena jaraknya 6 sepersepuluh inci lebih besar dari dimensi lebar dalam selungkup ideal dan beberapa perutean bantuan diperlukan untuk melengkapi pemasangan yang baik dan memastikan Segel udara yang bagus Tanduk dapat dipasang pada baffle 34 inci yang terbuat dari kayu yang sama, dan tidak memerlukan sisi atau kotak, hanya baffle depan dan beberapa cara untuk mendukungnya sudah cukup. Jika Anda seorang pemandu audio bersenandung emas, mungkin Anda ingin menjelaskan berapa waktu kedatangan sinyal akustik pada posisi mendengarkan Anda untuk melakukan ini, yang perlu Anda lakukan hanyalah memindahkan si tweeter kembali ke tanduk datar ke titik di mana Bagian belakang dua rakitan magnet itu berbaris vertikal, dan memindahkan seluruh rakitan ke depan hingga dalam jarak 3 inci dari posisi rakitan magnet pengemudi midrange. Jika Anda melakukan ini dan baffle atau dinding tanduk akhirnya membayangi permukaan pemasangan pengemudi midrange, cukup antarkan semua permukaan yang menghadap (yang memiliki pandangan pengemudi midrange) dengan Sonex atau suara serupa yang menyebar, busa yang tidak beraturan. Lakukan hal yang sama di atas tanduk 2382 sehingga tweeter tidak akan menyemprotkan suara ke permukaan yang memantulkan. MENGHUBUNGKAN SISTEM Setelah Anda selesai membuat lemari dan memasang semua driver dan tanduk dan melakukan semua yang Anda bisa untuk meredakan keluarga Anda sehingga Anda tidak memerlukan bantuan psikiatri rawat jalan, Anda dapat menghubungkan semuanya. Mulailah dengan membuat kabel speaker dari kawat terberat yang dapat Anda temukan - kabel baterai tidak terlalu besar. Keuntungan kabel kuotototeretik hanya memiliki kabel pengeras suara biasa adalah bahwa alat ini biasanya berukuran lebih berat, di luarnya tidak ada yang terukur (atau itu Pasti sudah diterbitkan) beda. Potong kabel Anda lebih lama dari yang Anda pikir akan Anda perlukan untuk menjalankan minimum, tapi hati-hati untuk mencari amplifier daya yang dekat dengan speaker sehingga tidak ada kabel ekstra. Catat semua kabel Anda dengan hati-hati (VLF, LF, MF, HF) untuk polaritas kiri dan kanan serta catat jika perlu sehingga Anda tidak bingung, dan menjadi sangat membantu, Anda seharusnya bisa merasakan tanda-tanda dalam kegelapan atau di belakang. Rak ampli jika Anda bekerja di ruang terbatas. Rak ampli (crossover dan power amplifier) harus dihubungkan sesuai dengan praktik rekayasa logis, penyambungan kabel sinyal dan speaker pada sudut siku-siku dan mengisolasi alasan chassis yang diperlukan untuk mencegah ground loop dan hum. Seharusnya merakit dan memasang rak ampli Anda sehingga tidak ada dengungan, hanya beberapa desis (terkait dengan pengeras suara sensitivitas tinggi) dari amplifier saat kontrol gain mereka terbuka lebar. Crossover 525 harus diatur untuk membagi driver subwoofer (VLF) dan midbass (LF) pada 100 Hz. Penguat daya 6290, pada gilirannya, terhubung ke dua pasang driver 18quot yang dihubungkan paralel ke masing-masing saluran, dan kedua 6260-an beralih ke mode mono yang dijembatani dan masing-masing mengemudikan salah satu driver midbass. Output MF dari feed masing-masing 525s masing-masing 6260, diatur ke mode mono dijembatani, yang dihubungkan pada gilirannya, ke driver kelas menengah. Frekuensi pemisah untuk driver LF-MF harus diatur ke 500 Hz. Keluaran HF dari 525s memberi makan pasangan (amplifier) 6230 power amplifier yang tersisa yang pada gilirannya memberi makan salah satu dari 3105 crossover pasif. Frekuensi pemisah bagian MF-HF harus diset ke 1200 Hz. Driver kompresi 2445J dihubungkan ke output frekuensi rendah masing-masing 3105, dan tweeter 2405 masing-masing terhubung ke output frekuensi tinggi dari 3105 crossover. Hook up driver midbass 15quot dalam polaritas terbalik dari driver 18quot. Kaitkan driver midrange dengan polaritas terbalik ke driver midbass (polaritas yang sama dengan driver 18quot). Tanduk dan tweeter, melalui 3105-an, harus dihubungkan sesuai instruksi merah-hitam pada lembar instruksi crossover 3105 dan dihubungkan sedemikian rupa sehingga masukan ke 3105 (terminal merah) adalah polaritas terbalik dari pengemudi midrange, kecuali jika Anda memiliki selaras secara fisik. Tanduk dan tweeter ke depan pada pengemudi midrange, dalam hal ini Anda harus membalik polaritas masukan 3105 itu. (CATATAN: Item yang satu ini mungkin memerlukan beberapa fudging dan penyesuaian termasuk eksperimen polaritas, untuk mencapai karakteristik penundaan kelompok terbaik). TUNING DAN TWEAKING Setelah Anda selesai menggabungkan semuanya dan menyiram semuanya secara profesional ke dinding ruang tamu Anda, Anda perlu mendapatkan penganalisis spektrum 13 oktaf atau insinyur audio yang memilikinya, dan mengatur semuanya dengan benar dengan mengatur kontrol kontrol dan sejenisnya. Jika Anda tinggal di area metropolitan, Anda mungkin menemukan seseorang dengan mesin TEF yang cukup penasaran untuk mengukur dan men-tweak sistem yang hampir tidak pernah dia lihat. Saya sarankan agar Anda tidak mencoba memainkan musik apapun melalui sistem sampai beberapa pengukuran dan penyesuaian dapat dilakukan, sehingga Anda tidak memiliki kesempatan untuk menderita penyesalan oleh pembeli saat, karena sistem tidak diatur dengan benar, tidak benar. Jika Anda menghabiskan banyak uang ini, Anda berhutang pada diri sendiri untuk menyelesaikan pekerjaan dengan benar. Prosedur terbaik untuk menentukan gain yang benar antara semua amplifier adalah dengan menggunakan pita noise pink yang difrekuensi tinggi. Jika filter pita oktaf tidak tersedia, gunakan aturan ibu jari bahwa subwoofer merupakan bagian yang paling tidak sensitif dari sistem, jadi mereka harus digunakan sebagai acuan tingkat untuk komponen lainnya, dengan kata lain, membalikkannya sepanjang jalan. , Kemudian muncul midbass, midrange, dan tanduk, dalam urutan itu, sampai tingkat terdengar seperti yang mereka padukan. Kemampuan pengukuran respons frekuensi pada sistem pengukuran TEF mungkin adalah cara terbaik untuk memastikan pengaturan sistem yang tepat dan kemampuan pengukuran energi dan fasa mesin memudahkan penyesuaian fisik komponen di sepanjang sumbu Z, maju atau mundur dengan tepat. TEORI OPERASI Filosofi saya pada desain sistem speaker sesuai dengan JBL. Secara sederhana, output daya akustik dari sistem speaker di lapangan yang berdifusi dan meragukan, harus sejujur mungkin. Masing-masing elemen pengemudi harus lebih kecil dari panjang gelombang yang diminta untuk diperbanyak. Saya juga merasa bahwa tidak ada elemen sistem yang harus ditekankan selama operasi pada tingkat mendengarkan yang khas. Untuk yang terakhir dan saya percaya alasan yang paling penting, saya telah memilih driver midbass dan midrange yang paling efisien yang tersedia untuk memulai dengan keuntungan operasi yang berada di bawah 1 persen dari kemampuan daya pengenal. Anda harus menemukan, saat mendengarkan sistem ini, ada kualitas sonik yang lebih besar daripada kehidupan yang menghasilkan reproduksi sinyal masukan yang sangat rinci dan mengungkap. Hal ini disebabkan oleh sensitivitas komponen sistem yang tinggi. Meskipun ada banyak alasan untuk menginginkan satu driver kecil untuk mereproduksi keseluruhan spektrum audio, kita tahu dari pengalaman langsung bahwa pengemudi kecil tidak dapat menangani daya yang cukup untuk menghasilkan output akustik yang cukup. Kerucut speaker 4 inci harus bisa bergerak maju mundur 4 kaki untuk menggerakkan udara sebanyak mungkin subwoofer dalam sistem ini mampu bergerak. Selain itu, semakin lebar rentang frekuensi yang harus ditutup oleh pengemudi, semakin banyak distorsi Doppler suara irama non-harmonis dan non-musik yang disebabkan oleh modulasi suara frekuensi tinggi yang disebabkan oleh gerakan diafragma besar yang terkait dengan frekuensi rendah simultan. reproduksi. Jawaban untuk distorsi Doppler dan kapasitas penanganan daya adalah membagi spektrum frekuensi audio menjadi band, yang masing-masing mewakili sebagian kecil dari daya yang dibutuhkan total dan masing-masing memerlukan penggerak yang lebih kecil secara berturut-turut untuk menyebarkan panjang gelombang yang secara berturut-turut lebih kecil dari pita frekuensi yang dibutuhkan. . Inti dari kinerja sistem adalah kemampuannya untuk melacak transien, yang, dalam perangkat lunak musik yang direkam dengan baik, akan memiliki tingkat puncak 20 sampai 30 desibel yang lebih tinggi daripada rata-rata daya yang digunakan untuk bermain pada tingkat mendengarkan yang wajar. Speaker efisiensi rendah menderita pemanasan koil suara mereka dan kompresi output berikutnya, dari input daya tinggi. Pemikiran saya adalah bahwa untuk pengeras suara dengan setia mereproduksi sinyal masuk, harus pada setiap saat, bertindak seolah-olah sinyal adalah stimulus pertama yang diterima, tidak mungkin loudspeaker akurat jika sinyal hanya direproduksi mengubah Pengeras suara karakteristik listrik atau mekanik, misalnya dengan memanaskan koil suara atau meregangkan bahan aktif yang membentuk bagian pengeras suara. Dalam kasus speaker elektrostatik, kerugian terjadi akibat kemampuan gerak dan daya yang terbatas. Speaker elektrostatik juga mengalami efisiensi yang sangat rendah. Solusinya adalah untuk menjaga tingkat daya input nominal nominal rendah sehingga pemanasan diminimalkan, dan untuk melakukan ini yang diperlukan untuk menggunakan efisiensi tinggi driver sebagai elemen sistem. Kerugian dari driver efisiensi tinggi adalah bahwa mereka meliput pita frekuensi yang lebih sempit seiring efisiensi mereka meningkat. Sebaliknya, pengemudi bandwidth lebar (JBL LE8 adalah sebuah contoh) selalu menunjukkan efisiensi rendah - manifestasi langsung dari hukum fisika. Anda mungkin bertanya-tanya mengapa diperlukan untuk menyediakan amplifier berkekuatan 600 watt untuk driver yang akan dioperasikan secara nominal pada watt. Puncak sementara musik desibel 20 desibel memerlukan 100 kali daya yang dibutuhkan oleh sinyal rata-rata dan puncak desibel 30 memerlukan 1000 daya yang dibutuhkan oleh sinyal rata-rata. Kemampuan output 600 watt penguat yang menggerakkan unit midbass hanya mewakili sedikit kurang dari 28 desibel di atas 1 watt catu daya untuk melacak transien. Jika Anda adalah penggemar speaker elektrostatik atau Bi-polar, Anda akan membenci suara sistem ini sampai Anda terbiasa, setelah itu Anda akan membenci jenis elektrostatik dan bi-polar. Analogi efek yang dirasakan adalah tipe sistem (tipe efisiensi tinggi) ini seperti mengeluarkan kompresor elektronik dari sistem speaker yang baik. Ada yang pasti menjadi quintime-smearingquot atau quotimage-smearingquot dari sumber suara manapun yang bukan titik sederhana di ruang angkasa, namun dengan menyelaraskan elemen sistem pada garis vertikal lurus (kecuali driver subwoofer), waktu horisontal dan tangkapan gambar dieliminasi. . Manusia tidak merasakan waktu vertikal dan noda gambar kecuali mereka melompat-lompat di depan sistem speaker - sebuah praktik yang tidak saya sarankan untuk mendengarkan secara kritis (membagi perhatian Anda). Karena pengeras suara komponen JBL masing-masing sangat sesuai dengan praktik manufaktur, pencitraan stereo dari sistem ini sangat spektakuler. DISCLAIMER: FIRMAN SERIUS PERINGATAN Sistem yang dijelaskan di sini dengan mudah mampu menghasilkan tingkat tekanan suara jauh di atas yang menyebabkan gangguan pendengaran ireversibel - jangan anggap enteng ini. Anda mungkin menderita tidak hanya gangguan pendengaran permanen, tapi juga dering konstan di telinga yang dapat menyebabkan insomnia dan menyebabkan gangguan saraf atau masalah emosional. JBL dan penulis ini tidak membuat klaim dan tidak bertanggung jawab atas disain, operasi atau konsekuensi penggunaan sistem yang dijelaskan di sini. 1988 Drew DanielsDistortion Dalam Power Amplifier. Diperbarui: 9 Okt 2001 Distorsi yang dihasilkan oleh power amplifier Kelas-B solid-state yang khas ditunjukkan terdiri dari delapan mekanisme, yang kesemuanya dapat ada dan produk distorsinya saling melengkapi untuk memberikan hasil yang kompleks. Metode untuk mengisolasi setiap mekanisme studi, dan meminimalkan kontribusinya, diberikan. Jika distorsi yang dapat dihindari dirancang, amplifier Kelas-B dengan distorsi yang sangat rendah (Di bawah 0.0005 pada 1 kHz, 0,003 pada 10 kHz) dapat dirancang sebagai masalah rutin, dan tanpa biaya tambahan yang signifikan. Amplifier semacam itu menentukan patokan distorsi, dan saya menamakannya amplifier tanpa ampun. KLIK DI BAWAH UNTUK MELIHAT DIRECT TO BAGIAN. KLIK PADA ANGKA UNTUK VERSI FULL-SIZE. C O N T E N T S 0: Pendahuluan. 1: Konfigurasi Amplifier Generik. 2: Delapan Distorsi. 3: Tiga Distorsi Tidak Ada. 4: Teknik Investigasi Amplifier. 5: Mekanisme Distorsi. 6: Konsep Amplifier Tak Bersuara. 7: Kesimpulan. Referensi. Jauh lebih detail tentang distorsi dan hal-hal lain dapat ditemukan dalam buku yang akhirnya saya tulis: 0. PENDAHULUAN. Mengingat pentingnya kekuatan penguat daya audio, secara mengejutkan sedikit informasi yang dapat dipercaya telah dipublikasikan mengenai desain mereka. Distorsi khususnya telah terbengkalai, meski merupakan fitur penguat performa paling bervariasi. Anda mungkin memiliki dua unit yang ditempatkan di sisi, satu memberi 2 THD dan 0,0005 lainnya dengan kekuatan penuh, dan keduanya mengklaim untuk memberikan pengalaman audio tertinggi. Saya menyelidiki asal mula distorsi pada periode 1992-94, dan menentukan bahwa distorsi penguat daya, yang secara tradisional merupakan hal yang sulit dan misterius untuk digali, adalah penggabungan delapan mekanisme dasar, dilapiskan dan kadang-kadang sebagian dibatalkan, memberikan hasil yang kompleks. Saya mengembangkan cara untuk mengukur dan meminimalkan setiap mekanisme distorsi secara terpisah, dan hasilnya adalah metodologi perancangan untuk membuat amplifier Kelas-B atau Kelas-A dengan kinerja distorsi yang begitu baik sehingga dua atau tiga tahun yang lalu hal itu dianggap tidak mungkin dilakukan. 0.0008 pada 1 kHz dan 0,003 pada 10 kHz mudah didapat. Metodologi ini memberikan hasil yang andal dan berulang dengan jumlah umpan balik negatif yang moderat meningkat dalam kompleksitas dan biaya tidak signifikan. 1. KONFIGURASI AMPLIFIER GENERIK. Gambar 1a menunjukkan sirkuit penguat daya Lin generik, dengan tahap masukan diferensial universal sekarang, mewakili sesuatu seperti 98 amplifier yang pernah dibuat. Ini adalah titik awal yang jelas untuk penyelidikan penguat. Gambar 3 menunjukkan plot distorsinya ada dua regu distorsi. THD di bawah 1 kHz rendah pada 0,002 namun tidak nol, lantai kebisingannya kira-kira 0,0006. Di atas 1 kHz, THD quadruples dengan masing-masing oktaf dan mencapai 0,5 sebelum 20 kHz. Topologi dasar adalah penguat transkonduktansi (input tegangan-beda, keluaran arus) yang menggerakkan konstanta Tegangan Tegangan (current-to-voltage converter) Tegangan Amplifier, diikuti oleh penyangga daya gain-satu. Tegangan pada basis transistor VAS biasanya hanya beberapa milivolt, dan sedikit diminati sendiri, arusnya beralih dari tahap masukan ke VAS yang diperhitungkan. Topologi ini memiliki banyak keunggulan, termasuk kompensasi sederhana. 2 Daftar Isi Bagian 3 2. DISTORSI DELAPAN. Mekanisme distorsi penguat daya generik terbagi dalam delapan kategori dasar. Distorsi 3 adalah yang dihasilkan oleh tahap keluaran, dibagi menjadi tiga mekanisme yang berbeda 3a, 3b, 3c yang tidak terkait dengan asal fisiknya. Demikian pula, Distorsi 8 biasanya hanya terjadi di kapasitor di bagian bawah lengan umpan balik, namun, dalam desain AC-coupled kapasitor output dapat menyebabkan distorsi yang signifikan. Klasifikasi ini mengasumsikan tidak ada pemindaian kliping, overload, slew-limiting, atau parasit. 1 Tahap input (seimbang) Dengan demikian THD pada 1,47 kali gain (.00404) bila diskalakan untuk keuntungan CL yang realistis sebesar 23, dikurangi dengan faktor (231,47) 2 245, memberikan nilai 0,000017 pada 15 kHz. 3.3 Distorsi Termal. Distorsi termal kadang-kadang digambarkan sebagai yang disebabkan oleh perubahan suhu siklik pada frekuensi sinyal, parameter perangkat modulasi. Ini adalah masalah nyata di IC, dengan perangkat input dan output di dekat kedekatan termal, namun dalam penguat daya komponen diskrit tidak ada penggabungan seperti itu, dan tidak ada distorsi semacam itu. Distorsi termal diharapkan muncul sebagai kenaikan distorsi harmonik kedua dan ketiga pada frekuensi sangat rendah, dan efek terbesarnya adalah pada tahap keluaran Kelas-B dimana disipasi sangat bervariasi selama satu siklus. Efeknya sama sekali tidak ada. Ini mungkin karena perangkat penggerak dan keluaran memiliki sambungan besar dengan inersia termal tinggi. Sebuah transistor driver MJE340 memiliki area chip empat kali dari TL072, jadi parameter seperti VBE agaknya tidak dapat banyak berubah bahkan pada 10 Hz. Faktor NFB global juga tertinggi di LF. Dengan menggunakan metodologi desain, sebuah penguat dapat didesain dengan mudah untuk menghasilkan kurang dari 0,0006 THD pada 10Hz (150W8-Ohm) tanpa mempertimbangkan distorsi termal ini menunjukkan bahwa ini bukan masalah. Plot THD yang meningkat pada frekuensi rendah sering terjadi, namun saya selalu menemukan kenaikan LF dapat dieliminasi dengan memperbaiki defoupling yang rusak (Distortion 5) atau meningkatkan kapasitor umpan balik. (Distorsi 8) Sebagai argumen lebih lanjut, perhatikan sisa distorsi penguat Class-B underbiased, dengan menggunakan keluaran CFP sehingga bias diam bergantung pada suhu pengemudi saja. Ketika tenaga sinewave dikirim ke sebuah beban, lonjakan crossover (yang dihasilkan oleh underbiasing) pada residu THD perlahan-lahan mengurangi tingginya selama beberapa menit saat driver melakukan pemanasan. Ketinggian paku ini memberi indikasi terus menerus suhu pengemudi, dan variasi yang lambat mengindikasikan konstanta waktu termal puluhan detik, dan respons yang tidak berarti pada 10Hz. 4.1 Gain Open-Loop dan Pengukurannya. Kinerja distorsi loop tertutup penguat adalah produk linearitas loop terbuka dan faktor umpan balik negatif. Untuk gain loop tertutup tetap, faktor NFB ditentukan oleh gain loop terbuka dan variasinya dengan frekuensi. Modifikasi rangkaian yang khas - misalnya mengubah nilai R2 pada Gambar 1 - mengubah gain loop terbuka serta linearitas dan penting untuk mengetahui apakah perubahan yang diamati disebabkan oleh peningkatan linieritas OL, atau peningkatan OL yang meningkat. Oleh karena itu, dibutuhkan metode pengukuran OL yang cepat dan mudah. Metode standar untuk gain loop terbuka op-amp melibatkan pemecahan umpan balik dan manipulasi gain loop tertutup (CL), prosedur yang tidak mungkin berhasil dengan penguat daya rata-rata. Untuk penguat generik pada Gambar 1, gain loop terbuka adalah tegangan keluaran yang dibagi oleh tegangan diferensial pada input. Jika respons frekuensi CL datar, sebidang gain loop terbuka versus frekuensi diperoleh dengan mengukur tegangan kesalahan di antara input, dan merujuknya ke tingkat keluaran. Ini memberikan plot terbalik naik di HF daripada jatuh, karena penguat membutuhkan lebih banyak voltase kesalahan untuk keluaran yang sama saat frekuensi meningkat. Gambar 2 menunjukkan penguatan OL dari penguat pada Gambar 1. Dengan adanya testgear dengan input seimbang CMRR yang tinggi, metode ini adalah penyangga sederhana masukan diferensial penguat dari kapasitansi kabel dengan pengikut tegangan TL072, yang menempatkan beban yang tidak dapat diabaikan pada sirkuit, dan mengukur tingkat Sehubungan dengan outputnya. CMRR testgear mendefinisikan gain loop terbuka maksimum yang dapat diukur. Sistem Audio Precision-1 bekerja sangat baik di sini. Sebuah plot kalibrasi (jejak bawah pada Gambar 2) dihasilkan dengan memberi makan dua masukan penyangga dari sinyal yang sama ini juga meningkat pada 6dBoctave, karena asimetri masukan testgear, dan setidaknya harus 10dB di bawah sinyal kesalahan penguat untuk akurasi. Kurva merata pada LF, dan mungkin naik, karena ketidakseimbangan kapasitor penghambat input testgear ini membuat penentuan kutub terendah P1 sulit, namun P1 bukanlah parameter vital tersendiri. 4.2 Amplifier Model. Bagian sinyal kecil linier adalah titik awal yang jelas untuk penguat distorsi distorsi rendah 1 dan 2 dapat dengan mudah mendominasi kinerja penguat dan perlu dipelajari tanpa komplikasi pada tahap keluaran Kelas B. Sirkuit ini direduksi menjadi penguat model yang terdiri dari tahap masukan dan VAS saja, ditambah kelas penguat emitor Kelas A yang sangat linier sebagai tahap keluaran untuk mendorong jaringan umpan balik tidak ada pemuatan eksternal. Model disini berarti mengurangi arus daripada tegangan. Penguat model harus mampu memberikan ayunan voltase daya penuh, karena distorsi pasangan input bergantung pada tingkat keluaran absolut, dan bukan proporsi tegangan rel yang dilalui. Amplifier model tanpa tahap keluaran yang lambat dapat memberikan hasil yang sangat optimis untuk stabilitas HF. Faktor tinggi NFB yang stabil dalam penguat model dapat dengan mudah menjadi tidak stabil saat tahap keluaran nyata ditambahkan. Nilai Cdom harus diantisipasi untuk penguat lengkap. Sebuah plot THD khas dari model amp seperti pada Gambar 1 meningkat dengan kemiringan yang curam, karena kenaikan awal pada 6dBoctave dari VAS berkontribusi, dan kemudian didominasi oleh, peningkatan distorsi 12dBoctave dari tahap input yang tidak seimbang. (Lihat 5.1.2) 4.3 simulasi SPICE. This is a powerful technique I use PSpice. SPICE gives insight into the open-loop linearity of both input and output stages, but applying it to the VAS is problematical as BJT Early Effect is implemented as a linear approximation. This seems unlikely to give accurate results for a stage with a large signal on its collector. Section 3 Contents Section 6 5. THE DISTORTION MECHANISMS. 5.1 DISTORTION 1. Input Pair Non-linearity The input differential pair implements one of the few forms of distortion cancellation that is truly reliable - the transconductance of the input pair are determined by transistor physics rather than matching of variable parameters such as beta. The logarithmic relation between Ic and Vbe is proverbially accurate over eight or nine decades of collector current. The prime motivation for using a differential pair as the input stage of an amplifier is its low DC offset. Apart from cancellation of the Vbe voltages, it has the extra advantage that the standing current does not flow through the feedback network. A second powerful reason, which seems less well-known, is that linearity is far superior to single-transistor input stages. Transconductance (gm) is maximal at Vin0, when the two collector currents are equal, and this maximum is proportional to the tail current Ie. 4 Device beta does not figure in the equation, and linearity of the input pair is not significantly affected by transistor type. The transconductance plot in Fig 4 shows the linearising effect of local feedback (emitter degeneration) on the voltage-incurrent-out law it plots transconductance against input voltage and demonstrates how emitter degeneration reduces peak transconductance, flattening the curve over a wider input range. Emitter degeneration markedly improves input stage linearity, but the overall amplifier NFB factor is reduced, for the vital HF closed-loop gain is determined solely by input-stage transconductance and the value of the dominant-pole capacitor. (Eqn 2) FETs seem a poor idea for the input stage. The basic gm is so low compared with BJTs that there is little scope for linearisation by adding source resistors for local degeneration, so an FET input stage will be very non-linear compared with a BJTs degenerated down to the same transconductance: see 5.1.3. Curve A in Fig 5 shows the distortion plot for a model amplifier, (5 Vrms output) designed so all distortion is negligible apart from that from the input stage with a class A output this simply means ensuring that the VAS is properly linearised. Note the vanishingly low LF distortion. For R2 10K, distortion is below the .001 noise floor until it emerges at 1 kHz, rising steeply at 12 dBoctave. This rapid increase is due to the input stage signal current doubling every octave, to feed Cdom therefore the associated second harmonic distortion doubles with each octave increase. Simultaneously the overall NFB available to linearise this distortion falls at 6dBoctave, and the combined effect is an quadrupling or 12 dBoctave rise. If the input stage is properly balanced, only third harmonic is generated, which quadruples rather than doubling as amplitude doubles, resulting in a 18 dBoctave slope however this only appears much further up the frequency range, and the total distortion produced is much less. If the VAS or output stage generates distortion it rises at only 6dBoctave, and looks quite different. 5.1.1 Input stage distortion in isolation. For serious research we need to measure input-stage non-linearity open-loop and in isolation. This is simply done with the test circuit of Fig 6. The current-to-voltage conversion op-amp uses shunt feedback to generate an AC virtual-earth at the input-pair output, and uses a third -30V rail to allow the ip pair collectors to work at a realistic DC voltage just above the V - rail the 10K feedback resistor may be scaled to prevent op-amp clipping. Input DC balance is set by the 10K pot the THD residual diminishes as balance is approached, until the second - harmonic is nulled, leaving almost pure third harmonic. 5.1.2 Input stage balance. Exact DC balance of the input differential pair is essential for minimum distortion. It seems almost unknown that even minor deviations from equality of collector current (Ic) in the input devices seriously upset the 2nd-harmonic cancellation, by moving the operating point from A to say, B, in Fig 4. The gm is both less and changing faster at B, so imbalance reduces open-loop gain as well as increasing distortion. The effect of small amounts of Ic imbalance is shown in Fig 7 Table 3 with an input of -45dBu an Ic imbalance of only 2 seriously worsens linearity, THD increasing from 0.10 to 0.16, while for 10 imbalance this deteriorates to 0.55. Ic balance needs an accuracy of 1 or better for lowest distortion at HF, where the input pair works hardest. Imbalance in either direction gives similar results. This explains the complex distortion changes that accompany the apparently simple experiment of altering the value of R2. We might design an input stage as in Fig 8a, where R1 has been selected as 1K by uninspired guesswork and R2 made high at 10K in a plausible but misguided attempt to maximise OL gain by minimising TR2 collector loading. R3 is also 10K to give the stage a notional balance unhappily this is a visual rather than electrical balance. The asymmetry is shown in the resulting collector currents this design generates a lot of avoidable second harmonic distortion, displayed in the 10K curve of Fig 5. Recognising the crucial importance of Ic balance, the circuit can be rethought as Fig 8b. If the collector currents are to be equal, R2 must be twice R1, as both have about 0.6V across them. The dramatic effect of this simple change is shown in the 2K2 curve of Fig 5 the improvement is accentuated as OL gain has also increased by some 7 dB, though this has only a minor effect on the closed-loop linearity compared with the improved input stage balance. R3 has been removed as it contributes nothing to input balance. The input pair can be approximately balanced by the correct values for R1 and R2, but we remain at the mercy of several circuit tolerances. The current-mirror configuration in Fig 8c forces the two collector currents very close to equality when global NFB is applied, giving excellent cancellation of the second harmonic the great improvement is seen in the current-mirror curve of Fig 5. A simple mirror has well - known Ib errors but they are not large enough to affect distortion. The hyperbolic-tangent law also holds for the mirrored pair, 5 but the output current swing is twice as great for the same input voltage. This doubled output is at the same distortion as a perfectly-balanced non-mirror input, as linearity depends on the input voltage, which has not changed. Putting a current-mirror in a well-balanced input stage therefore increases the total OL gain by at least 6dB, and possibly by up to 15dB if the stage was previously poorly balanced the compensation by Cdom must allow for this. Another happy consequence is that slew-rate is roughly doubled, as the input stage can now source and sink current into Cdom without wasting some in resistive collector load R2. If Cdom is 100pF, the slew-rate of Fig 9a is about 2.8Vusec up and down, while 9b gives 5.6Vusec. The unbalanced pair at Fig 8a displays further vices by giving 0.7Vusec positive-going and 5Vusec negative-going. A discrete current-mirror needs its own emitter-degeneration for accuracy. A voltage-drop across the mirror emitter-resistors of 60mV is enough to make the effect of Vbe tolerances negligible without degeneration there is significant variation in HF THD with different transistor specimens. To summarise, the advantages of a mirrored input stage are that second-harmonic distortion is eliminated, and maximum slew-rate is doubled. 5.1.3 Improving input-stage linearity. Even if the input pair has a current-mirror, HF distortion can still be excessive once it emerges from the noise floor it octuples with each doubling of frequency, and so it is well worth postponing the evil day until as far as possible up the frequency range. Input stage transconductance increases with Ic, so it is possible to raise gm by increasing the tail-current, and then reduce it back to its previous value (otherwise Cdom must be increased to maintain stability) by applying local NFB in the form of emitter-degeneration. This greatly improves input linearity, despite its rather unsettling flavour of something-for-nothing. Input transistor non-linearity can be regarded as an internal non-linear emitter resistance re, and we have reduced the value of this (by increasing Ic) and then replaced the missing part with a linear external resistor Re. The original input stage in Fig 1 has a per-device Ic of 600uA, giving a differential (ie, mirrored) gm of 23 mAV and re 41.6 Ohm. The improved version in Fig 9b has Ic 1.35mA and so re 18.6 Ohm emitter degeneration resistors of 22 Ohm are added to reduce gm back to its original value, as 18.6 22 is approx 41.6 Ohm. The THD measured by the circuit of Fig 6 for a -40dBu input voltage falls from 0.32 to 0.032, an extremely valuable linearisation which translates into an HF distortion reduction of about 5 times for a complete Class-B amplifier the full advantage is rarely gained. The remaining distortion is still visually pure third-harmonic if the input pair is balanced. The reduction of re is limited by the need for practical values of tail current. As a further benefit, increasing the tail current also increases slew rate. 5.2 DISTORTION 2 The Voltage-Amplifier Stage (or VAS) is often regarded as a critical part of a power-amplifier. It provides all the voltage gain and simultaneously the full output voltage swing. However, as is not uncommon in audio, all is not quite as it appears. A well-designed VAS stage contributes relatively little to the total distortion of an amplifier if even the simplest steps are taken to linearise it further, its contribution disappears. This is because the action of Miller dominant-pole compensation in this stage is rather elegant. It is not simply a matter of finding the most vulnerable transistor and setting it in treacle. As frequency rises and Cdom takes effect, negative feedback is no longer applied globally around the whole amplifier, which would include the higher poles, but instead is smoothly transferred to a purely local role in linearising the VAS. Since this stage is effectively a single transistor, a large amount of local NFB can be applied to it without stability problems. VAS distortion arises from the fact that the transfer characteristic of a common-emitter amplifier is curved, being a portion of an exponential. 6 This generates predominantly second-harmonic distortion, which in a closed-loop amplifier will increase at 6dBoctave with frequency. VAS distortion does not worsen for more powerful amplifiers because the stage traverses a constant proportion of its characteristic as the supply-rails are increased. This is not true of the input stage increasing output swing increases the demands on the transconductance amp as the current to drive Cdom increases. 5.2.1 Measuring VAS distortion in isolation. Isolating the VAS distortion for study requires the input pair to be specially linearised, to prevent its steeply-rising distortion characteristic from swamping the VAS contribution. This is done by heavily degenerating the input stage this also reduces open-loop gain, and the reduced global NFB factor exposes VAS non-linearity. See Fig 10, where the 6dBoctave slopes suggest an origin in the VAS. Distortion increases with frequency as Cdom rolls-off the global NFB factor. To confirm that this distortion is due solely to the VAS, it is necessary to find a method for experimentally varying VAS linearity while leaving all other circuit parameters unchanged. In a model amplifier this can be done simply by varying the V - voltage this varies the proportion of its characteristic over which the VAS swings, and thus only alters the effective VAS linearity, as input stage operation is not significantly affected. (Fig 10) The Vce of the input devices varies, but this has negligible effect. 5.2.2 VAS configurations. Various kinds of VAS are shown in Fig 11. It is important that the local open-loop gain of the VAS (that inside the local feedback loop closed by Cdom) be high, to linearise the VAS. Therefore a simple resistive collector load is unusable. Increasing the value of a resistive load to increase voltage gain decreases the VAS transistor Ic, reducing its gm and getting you back where you started. Local loop gain is enhanced by using an active load to increase the VAS collector impedance and thus increase the raw voltage gain either bootstrapping or a current-source do this effectively, though the current source is the usual choice. Both active-load techniques have another important role ensuring that the VAS can source enough current to drive the upper half of the output stage. If the VAS collector load was just a resistor to V, this capability would be lacking. The popular current source VAS is shown in Fig 11a. This works well, though the collector impedance and hence gain is limited by Early Effect and output stage loading. It is often stated that this topology provides current-drive to the output stage this is not really true. Once the local NFB loop has been closed by adding Cdom the impedance at the VAS output falls at 6dBoctave for frequencies above P1. With typical values the impedance is only a few kohm at 10kHz, and this hardly qualifies as current-drive. Fig 11b shows the bootstrapped equivalent. One drawback is that the increase in voltage gain is determined by the exact gain of the output stage, which is below unity and varies with loading. A more dependable form of bootstrapping is available if the amplifier incorporates a unity-gain buffer between the VAS collector and the output stage this is shown in Fig 11f, where R is the VAS collector load, defining VAS collector current by establishing the Vbe of the buffer transistor across itself. This voltage is constant, so R is bootstrapped and appears to the VAS collector as a constant-current source. A VAS current of 3mA is sufficient, compared with 6mA for the buffer stage. 5.2.3 VAS enhancements. The VAS distortion in Fig 10 shows the need for further improvement over that given by local NFB through Cdom, if our small-signal stages are to be distortion-free. The virtuous approach might be to try to straighten out the curved VAS characteristic, but in practice the simplest method is to increase the amount of local negative feedback around the VAS through Cdom. Equation 1 shows that LF OL gain (also the gain before Cdom is connected) is the product of input stage transconductance, TR4 beta and the collector impedance Rc. The last two factors represent the VAS gain and the local NFB can be augmented by increasing either. So long as Cdom remains the same, the global feedback factor at HF is unchanged and so stability is not affected. The effective beta of the VAS can be substantially increased by adding an emitter-follower. (Fig 11c) Adding an extra stage requires thought, for if additional phase-shift is introduced, the global loop stability will suffer. Here the extra stage is inside the Cdom Miller - loop and so there is little likelihood of trouble from this. The function of such an emitter-follower is sometimes described as buffering the input stage from the VAS but this is quite wrong its true function is VAS linearisation by enhancing local NFB through Cdom. Alternatively the VAS collector impedance can be further increased to get more local gain. This can be done with a cascode configuration - (see Fig 11d) but this technique is only useful when the VAS is not loaded by a seriously non-linear impedance. such as the input of a Class-B output stage. See section 5.4. The non-linear loading renders cascoding largely cosmetic unless a Class-A stage buffers the VAS collector from the output stage, as in Fig 11e. When a VAS-buffer is added, the drop in distortion is dramatic, as it is for the beta-enhancement method. The gain increase is ultimately limited by Early effect in the cascode and current-source transistors, and more seriously by the loading effect of the next stage, but it is of the order of 10 times and gives a useful effect. Fig 12 plots the distortion of a model amplifier with 100 Ohm input pair degeneration resistors, showing the extra distortion from a simple VAS. However, the beta-enhanced version has the THD submerged in the noise floor for most of the audio band, being well below 0.001. I think this justifies my contention that input-stage and VAS distortions need not be problems we have all but eliminated Distortions 1 and 2 from the list of eight. The beta-enhancing emitter-follower is slightly simpler than the buffered-cascode, but the cost difference is tiny. When wrestling with these kind of financial decisions it is as well to remember that the small-signal section of an amplifier usually represents less than 1 of the total cost, including mains transformer and heatsinks. Although the two VAS-linearising approaches look very different, the basic strategy of increased local feedback through Cdom is the same. Either method linearises the VAS into invisibility. 5.3 DISTORTION 3. The almost universal choice in semiconductor power amplifiers is a unity-gain output stage, specifically a voltage-follower. The most common output stages are shown in Fig 13 two versions of the double - emitter-follower, (EF) the Complementary Feedback Pair (CFP), and a source-follower FET output. The use of power FETs in output stages is often advocated. However, after much investigation, I have found the conclusion inescapable that FETs suffer not only from poor basic linearity, due to low gm, but also a crossover region that is inherently more jagged than BJTs. It is not possible to explore this in detail here, but see 7,8 A fundamental factor in determining output-stage distortion is the Class of operation. Apart from its inherent inefficiency, Class-A is ideal, having no crossover or switchoff distortion. Distortions 4, 5, 6 and 7 are direct results of Class-B operation and also disappear from a Class-A design. Distortion 1 (input-stage), Distortion 2 (VAS), and Distortion 3a (output-stage large-signal non-linearity) remain, however. Of those Class-A designs which have been published or reviewed, it is notable that the distortion produced is still significant. This need not be so see 9 for a Blameless amplifier biased into Class A, giving THD below 0.002, 10 Hz-20 kHz. It is not generally appreciated that moving into Class-AB, by increasing the quiescent current, does NOT simply trade efficiency for linearity. If the output power is above the level at which Class-A operation can be sustained, THD increases as the bias advances into AB operation. This is due to so-called gm-doubling (ie the voltage-gain increase caused by both devices conducting simultaneously in the centre of the output-voltage range, in the Class-A region) putting edges into the distortion residual that generate high-order harmonics much as under-biasing does. This vital fact is little known, presumably because gm-doubling distortion is at a relatively low level and is obscured in most amplifiers by other distortions. This is demonstrated in Fig 14a, b,c showing THD residuals for under - biasing, optimal, and over-biasing of a 150W8-Ohm amplifier at 1kHz. All non-linearities except Distortion 3 (output stage) have been eliminated. The over-biased case had its quiescent current increased until the gm-doubling edges in the residual had a 1:3 markspace ratio, and so was in Class A about one quarter of the time. All three traces were averaged 64 times to reduce noise the distortion in 14b is normally invisible in a 80 kHz measurement bandwidth. The RMS THD reading for Fig 14a was 0.00151, for 14b 0.00103, and for 14c 0.00153 Spectrum analysis of Fig 14c shows the higher harmonics to be at least 10dB greater than those for the optimal Class-B case, and comparable with 14a. In short, Class-AB offers lower distortion than Class-B below the AB threshold but more above it. Distortion 3a is the Large-Signal Non-linearity (LSN) that is produced by in both Class-A and B output stages, ultimately because of the large current swings in the active devices in bipolars, but not FETs, large collector currents reduce beta, leading to drooping gain at large output excursions. It excludes crossover and switchoff phenomena. Distortion 3b is classic crossover distortion, resulting from the non-conjugate nature of the two output halves. Distortion 3c is switchoff distortion, generated by the output devices failing to turn off quickly and cleanly at high frequencies, and is strongly frequency-dependent. The contributions of 3b and 3c to Distortion 3 occur in Class-B only. The linearity of the open-loop output stages in Fig 13 with typical values are shown in Figs 15,16,17. These diagrams were generated by SPICE, plotting incremental output gain against output voltage, with load resistance stepped from 16 to 2 Ohms, which I hope is the lowest impedance that feckless loudspeaker designers will throw at us. These plots have come to be known as wingspread diagrams, from their birdlike appearance. The power devices were Motorola MJ802 and MJ4502, which are more complementary than many so-called pairs, and minimise distracting large-signal asymmetry. The quiescent conditions are in each case set to minimise peak-to-peak gain deviations in the crossover region for 8-Ohm loading. The EF output stage. I have deliberately called this the Emitter-Follower (EF) rather than Darlington configuration, the latter implying an integrated device with driver, output, etc in one ill-conceived package. In the EF topology the input is transferred to the output via two base-emitter junctions in series, with 100 voltage feedback applied to each device separately to create cascaded emitter-followers. Fig 13a shows the most prevalent version (Type I) with driver emitter resistors R1,2 connected to the output rail. Type II uses one shared resistor Rd, and this improves HF switchoff (Fig 13b) basic linearity is the same, see Fig 15. The crossover region width is approx 10 V, and optimal bias 2.86 V..The CFP output stage. The other major type of bipolar complementary output is the Complementary Feedback Pair (CFP) or Sziklai Pair, seen in Fig 13c. The drivers now compare the output voltage with that at the stage input. Wrapping the outputs in a local NFB loop gives better linearity than EF versions with 100 feedback applied separately to driver and output transistors. The CFP topology is generally considered to show better thermal stability than the EF, because the Vbe of the output devices is inside the local NFB loop, and only the driver Vbe affects the quiescent conditions. The true situation is rather more complex. 10,11,12 The output gain plot is shown in Fig 16 Fourier analysis shows the CFP generates less than half the LSN of an emitter-follower stage. (See Table 4) It is hard to see why this topology is not more popular. The crossover region is much narrower, at about 1V. When under - biased, this appears on the distortion residual as narrower spikes than those from an emitter-follower output. Optimal bias here is 1.296V. Combining one of these stages with a distortionless small-signal section, and applying 30 dB of global NFB, we might expect an amplifier with vanishingly small THD. In fact, crossover distortion remains at HF, due to the difficulty of linearising high-order distortion with feedback that reduces with frequency Fig 18 shows the typical Blameless performance. 5.3.1 Large-Signal Nonlinearity. (Distortion 3a) LSN increases as load impedance decreases. In a typical output stage loaded with 8 Ohms or more, closed-loop LSN is negligible, the THD residual being almost entirely high-order crossover artifacts that are reduced less by NFB. As load impedance falls below 8 Ohms, third - harmonic appears in the residual, and soon dominates. The BJT output gain plots reveal that LSN is compressive, ie voltage gain falls with higher outputs. The fundamental reason for this gain-droop is the fall in output - transistor beta as Ic increases. 13 In the Emitter-Follower (EF) topology, beta falloff draws more output-base current from the driver emitter, pulling driver gain down further from unity this is the change in gain that affects the overall transfer ratio. Output-device gain is not directly affected, as given zero source impedance, beta does not appear in the equation for emitter-follower gain. As further evidence: In SPICE simulation, driving the output bases directly from zero - impedance voltage-sources (rather than drivers) abolishes the gain droop effect. The cause is in the output devices, but the effect is in the drivers. The SPICE Gummel-Poon model can be altered so output device beta does not drop with Ic (increase parameter IKF) and once more gain-droop does not occur, with drivers present. Measured LSN levels correlate well with the degree of beta-falloff shown in manufacturers data sheets. This holds for many different BJTs produced over the last 30 years. LSN does not appear to afflict FET outputs, which have no equivalent beta-falloff mechanism. See Fig 17 where the wings of the FET gain plot do not turn downwards at large outputs. LSN may be reduced in two ways: Use output devices that sustain beta well as Ic increases. The 2SC3281 and 2SA1302 transistors (Toshiba, Motorola) show much less beta-droop than average, and 4-Ohm distortion is reduced by about 1.4 times. Use two or more output devices in parallel even though this is unnecessary for handling the power output. Falloff of beta depends on collector current, and if two output devices are connected in parallel, the collector current divides in two between them, and beta-droop is much reduced. Doubling devices reduces distortion by about 1.9 times. These two techniques may be combined by using double sustained-beta devices. Doubled device results are shown in Fig 19 distortion at 80W4 Ohm has halved from 0.009 to 0.0045. 8 and 4 Ohm traces are now very close, the 4 Ohm THD being only 1.2 times higher. 5.3.2 Crossover distortion. (Distortion 3b) In a field like Audio where consensus of any sort is rare, it is widely acknowledged that crossover distortion is the worst problem afflicting Class-B power amplifiers. The pernicious nature of crossover distortion is that it occurs over a small part of the transfer characteristic, and so generates high-order harmonics. Worse still, this range is around the zero-crossing, so it is present at all levels, the THD percentage potentially increasing as output level falls, threatening very poor linearity at low powers. I investigated crossover distortion to see if it really did increase with decreasing output level in a Blameless amplifier. One problem is that an optimally-biased Blameless amplifier has such a low level of distortion at 1 kHz (0.001 or less) that the crossover artifacts are barely visible in circuit noise, even if low-noise techniques are used. Thus the THD percentage of the noise-plus-distortion residual is bound to rise with falling output, for the noise contribution remains constant this is the lowest line in Fig 20. To circumvent this, the amplifier was deliberately underbiased by varying amounts to generate ample crossover spikes these upper traces also rise as level falls, but Fig 20 shows that the THD percentage increases more slowly as level falls. Both EF and CFP output stages give similar results whatever the degree of underbias, THD increases by about 1.6 times as the output voltage is halved. In other words, reducing the output power from 25 W to 250 mW, which is pretty drastic, only increases THD by six times, and there is no sign of it increasing uncontrollably at low levels. Distortion versus level was also investigated at high frequencies, ie above 1 kHz where there is more THD to measure and optimal biasing can be used. Fig 21 shows THD versus level for the EF stage at a selection of frequencies Fig 22 shows the same for the CFP. Neither shows a sudden rise in percentage THD with falling level, though it is noticeable that the EF gives a good deal less distortion at lower power levels around 1 W. This is an unexpected observation, and is probably due to the greater width of the EF crossover region. To further get the measure of the problem, Fig 23 shows how HF distortion is greatly reduced by increasing the load resistance, providing further confirmation that almost all the 8 Ohm distortion originates as crossover in the output stage. The amount of crossover distortion produced depends crucially on optimal quiescent adjustment, so the thermal compensation used to stabilise this against changes in ambient temperature and power dissipation must be accurate. Investigation shows that the critical parameter is not quiescent current as such, but rather Vq, the quiescent voltage between the output device emitters see Fig 13. In both EF and CFP output stages, changing Re from 0.1 to 0.47 Ohms alters the optimal Iq considerably, but the values of Vbias and Vq barely change. Thus the voltage across the transistor base-emitter junctions and Res is what counts, not the resulting Iq. Selecting Re 0R1 for maximum efficiency is probably the over-riding consideration. This has the additional benefit that if the stage is erroneously over-biased into Class AB, the resulting gm-doubling distortion will only be half as bad as if the more usual 0R22 values had been used for Re. 5.3.3 Switchoff distortion. (Distortion 3c) This depends on the speed characteristics of the output devices and on the output topology. For topologies, the critical factor is whether the output stage can reverse-bias the output device base-emitter junctions to maximise the speed at which carriers are swept out, so the device is turned off quickly. The only conventional configuration that can reverse-bias the output junctions is the EF Type II, described below. The EF Type II configuration in Fig 13b is at first sight merely a pointless variation on Type I, but its valuable property is that the shared driver emitter-resistor Rd, with no output-rail connection, allows the drivers to reverse-bias the base-emitter junction of the output device being turned off. Assume that the output voltage is heading downwards through the crossover region the current through Re1 has dropped to zero, but that through Re2 is increasing, giving a voltage-drop across it, so TR4 base is caused to go more negative to get the output to the right voltage. This negative excursion is coupled to TR3 base through Rd, and can reverse bias it by up to -0.5V at 8 Ohms, increasing to -1.6V at 4-Ohms. Speed-up capacitor Cs improves this action, preventing the charge-suckout rate being limited by the resistance of Rd. A 1 uF speedup capacitor can half the THD at 40kHz, implying cleaner switchoff. The EF Type I has a similar voltage drop across Re2, but the connection of R1,R2 to the output rail prevents this from reaching TR3 base instead TR1 base is reverse-biased as the output moves negative. Charge-storage in the drivers is usually not a problem, so this does little good. Likewise, a CFP stage can only reverse-bias the driver bases, and not the outputs. The second influence on turnoff is the value of the driver emitter or collector resistors the lower they are the faster the stored charge can be removed. Applying these two criteria can reduce HF distortion markedly, but of equal importance is that it minimises overlap of output conduction at HF, which if unchecked gives an inefficient and potentially destructive increase in supply current. 14 5.4 DISTORTION 4: VAS loading distortion. Distortion 4 is that which results from the loading of the Voltage Amplifier Stage (VAS) by the non-linear input impedance of the Class-B output stage. The VAS collector impedance tends to be high, rendering it vulnerable to non-linear loading unless buffered or otherwise protected. The VAS is routinely (though usually unknowingly) linearised by applying local negative-feedback via the dominant-pole Miller capacitor, and this is a powerful argument against any other form of compensation. If VAS distortion still adds significantly to the amplifier total, then the local open-loop gain of the VAS stage can be raised to increase the local feedback factor. The obvious method is to raise the impedance at the VAS collector, and so the gain, by cascoding. However, if this is done without buffering the VAS, the loading will render the cascoding almost completely ineffective. A VAS-buffer eliminates this problem. The VAS collector impedance, while high at LF compared with other circuit nodes, falls with frequency as Cdom takes effect, so Distortion 4 is usually only visible at LF. It is also often masked by the increase in output stage distortion above dominant-pole frequency P1 as the amount of global NFB reduces. The fall in VAS impedance with frequency is demonstrated in Fig 24, obtained from a SPICE conceptual model.15 The LF impedance is that of the VAS collector resistance, but halves with each octave above P1. By 3 kHz the impedance is down to 1Kohm, and still falling. Nevertheless, it remains high enough for the input impedance of a Class-B output stage to significantly degrade linearity, the effect being shown in Fig 25. In 16 it was shown that as an alternative to cascoding, the VAS may be effectively linearised by adding an emitter-follower within the VAS local feedback loop, increasing the local NFB factor by raising effective beta rather than the collector impedance. As well as good VAS linearity, this establishes a much lower VAS collector impedance across the audio band, and is much more resistant to Distortion 4 than the cascode version. VAS buffering is not required, so this method has a lower component count. The only drawback is a greater tendency to parasitics near negative clipping, when used with a CFP output stage. Fig 26 confirms that the input impedance of an optimally-biased EF Type I output stage is highly non-linear even with an undemanding 8-Ohm load, the impedance varies by 10:1 over the output voltage swing. The Type II EF output has a 50 higher impedance around crossover, but the variation ratio is greater. CFP output stages have a more complex variation including a steep drop to below 20 KOhm around the crossover region. 5.5 DISTORTION 5: Decoupling errors. Most amplifiers incorporate small electrolytics (10 - 220uF) between each rail and ground to ensure HF stability. As a result rail-voltage variations cause current to flow into the ground. If an unregulated power supply is used, (and there are almost overwhelming reasons for doing so) the rails have non-zero AC impedance and bear voltage variations due to amplifier load currents as well as 100Hz ripple. In Class-B, the supply-rail currents are halfwave - rectified sine pulses, and if they contaminate the signal then distortion is badly degraded. The usual route for intrusion is via decoupling grounds shared with input or feedback networks, and a separate decoupler ground back to the star point is usually a complete cure.(Note that the star-point should be defined on a short spur from the heavy connection joining the reservoirs using B as the star point introduces hum due to the large reservoir-charging current pulses passing through it) Fig 27 shows the effect on an otherwise Blameless amplifier handling 60W8-Ohm, with 220uF rail decouplers at 1kHz distortion has increased by more than ten times, which is quite bad enough. However, at 20Hz the THD has increased 100-fold, turning a very good amplifier into a profoundly mediocre one by one misconceived connection. 5.6 DISTORTION 6: Induction from supply rails. Like Distortion 5, this stems directly from the Class-B nature of the output stage. The supply-rail currents are halfwave-rectified sine pulses, which can readily crosstalk into sensitive parts of the circuit by induction. This is very damaging to the distortion performance Fig 28 shows a large extra distortion component rising at about 6dBoctave. The distortion may intrude into the input circuitry, the feedback path, or even the output cables. This inductive effect was first publicised by Cherry 17. though the effect has been recognised by some practitioners for many years.18 This effect, apparently unfamiliar to most designers, seems to be a widespread cause of unnecessary distortion. The contribution of Distortion 6 can be reduced below the noise floor. Firstly, rigorously minimise loop areas in the input and feedback circuitry, ie keep each signal line very close to its ground. Secondly, limit the ability of the supply wiring to establish magnetic fields in the first place, by minimising the area of circuit loops carrying half - wave pulses. 5.7 DISTORTION 7: NFB Takeoff point distortion. There is a subtle trap in applying global NFB. Class-B output stages are awash with large halfwave-rectified currents, and if the feedback takeoff point is in slightly the wrong place, these currents contaminate the feedback signal, making it an inaccurate representation of the output voltage, and so introducing distortion Fig 29 shows the problem. At these current levels, all wires and PCB tracks must be treated as resistances, and it follows that point C is not at the same potential as point D whenever TR2 conducts. If feedback is taken from D, then a clean signal is established here, but the signal at output point C has a half - wave rectified sinewave added to it, due to the resistance C-D. The output will be distorted but the feedback loop does nothing about it as it does not know about the error. Fig 30 shows the practical result for an amplifier driving 100W into 8-Ohm. The resistive path C-D that did the damage was a mere 6mm length of heavy-gauge wirewound resistor lead. To eliminate this distortion is easy, once you are alert to the danger. Taking the NFB feed from D is not advisable as D is not a mathematical point, but has a physical extent, inside which the current distribution is unknown. Point E on the output line is much better, as half-wave currents do not flow through this arm of the circuit. 5.8 DISTORTION 8: Capacitor distortion. It seems to be little-known that electrolytic capacitors generate distortion when they have a significant AC voltage across them. It is even less well known that non-electrolytics show a similar effect in applications like Sallen Key high-pass filters. This has nothing to do with Subjectivist hypotheses about mysterious non-measurable effects this is all too real. Electrolytic distortion usually arises in DC blocking circuitry with significant resistive loading. Fig 31 shows the distortion for a 47uF 25V capacitor driving 8 Vrms into a 680 Ohm load. The distortion is a mixture of second and third harmonic, rising rapidly as frequency falls, at something between 12 and 18 dBoctave. The great danger of this mechanism is that serious distortion begins while the response roll-off has barely begun here THD reaches 0.01 at 20 Hz when the response is only down 0.2 dB. The voltage across the capacitor is 2.6 Volts peak, and this is a better warning of danger than the amount of roll-off. THD roughly triples as the applied voltage doubles the factor varies with capacitor voltage rating. The mechanism by which capacitors generate this distortion is unclear. Dielectric absorption appears to be ruled out as this is invariably modelled by adding linear components to the basic capacitor. Reverse biasing is not the problem, for DC biasing by up to 15V shows increased, not reduced distortion. Non-polarised electrolytics show the same effect but at a much greater AC voltage, typically giving the same distortion at one-tenth the frequency of a conventional capacitor of the same value cost and size generally rules out their use to combat this effect. The best solution is simply to increase the capacitor value until the LF distortion remains flat to 10 Hz. A small roll-off in the audio band is not a sufficient criterion. While the bandwidth of a system must be defined, using electrolytics in high-pass filters is never good design practice, because the tolerances are so large it is now clear they generate distortion as well. Capacitor distortion in DC-coupled power amplifiers is most likely to occur in the feedback network blocking capacitor. (C2 in Fig 1) The input capacitor C1 usually feeds a high impedance, but the feedback arm must have low resistances to minimise noise and DC offset. The feedback capacitor is thus an electrolytic, and if not quite large enough the THD shows a characteristic LF rise. Such LF rises are common, but need never occur. Capacitor distortion is usually the reason, but Distortion 5 (Rail Decoupling Distortion) can also contribute. They can be distinguished because Distortion 5 typically rises by only 6 dBoctave as frequency decreases, rather than 12 - 18 dBoctave for capacitor distortion. The distortion generated by an AC-coupled amplifiers output capacitor is more serious, as it is not confined to low frequencies. A 6800uF output capacitor driving 40 W into an 8-Ohm load gives mid-band third-harmonic distortion at .0025, as shown in Fig 32. This is five times more than a Blameless amplifier generates mid-band. Also, the LF THD rise is much steeper than in the small-signal case. 6: THE BLAMELESS AMPLIFIER CONCEPT. The basis of the design methodology is really the old clich Make the amplifier as linear as possible before applying Negative Feedback. In 5.1 and 5.2 it was demonstrated that the distortion from the small-signal stages can be made negligible compared with output-stage distortion, by balancing the input pair and adding local negative feedback to input and VAS stages. Likewise, 5.4 - 5.8 showed that Distortions 4 to 8 can be effectively eliminated by little-known but straightforward layout precautions. This leaves Distortion 3, in its three components, as the only distortion that is in any sense unavoidable, as Class-B stages free from crossover artifacts are so far beyond us. This leads to the concept of what I have called a Blameless Amplifier, the name being chosen to emphasise that the remarkably low THD comes from the avoidance of errors rather than from fundamental advances in circuitry. A Blameless Amplifier gives a distortion benchmark that varies relatively little if confined to 8-Ohm loading. It forms a well-defined point of departure for more ambitious and radical amplifier designs. So far I have used it as a basis for an extremely linear Class-A design 9. a Trimodal amplifier (so-called as it operates in any of the modes A, AB and B, as required) 19. and a Load-Invariant amplifier that minimises the THD increase with sub-8 Ohm loads.20 Above: Fig 33 shows the circuit of a Blameless Class-B amplifier. CLICK ON PICTURE FOR HIGHER-RES VIEW. Note that Fig 33 is only slightly more complex than the standard amplifier in Fig 1. The input pair now has a current-mirror to ensure input balance, and has undergone constant-gm degeneration, running at about 3.5 times the tail current of Fig 1. The VAS is linearised by addition of beta-enhancer TR12, and the remaining topological distortions were eliminated by careful layout. Performance is shown in Fig 18. I am aware that the distortion figures given here are unusually low for power amplifiers, but I would emphasise they are not freak results nor dependant on component selection. The only aspect of the linearity directly affected by device characteristics is distortion below 8 Ohms, as described in 5.3.1. So far more than twenty thousand 260W8-Ohm amplifiers based on the Blameless methodology have been built, with completely repeatable performance. 7. CONCLUSIONS. In this paper I have attempted a concise but complete account of power amplifier distortion. The linearity obtainable with relatively conventional circuitry is far better than one would suspect. It also shows that if power amplifier distortion is to be eradicated entirely, future work must be focused on the output stage distortions. DIAGRAM CAPTIONS. Fig 1: Fig 1a is the genericLin power amplifier circuit, with typical component values. 1b shows the small-signal Class-A output stage that replaces TR6-9 to make a model amplifier. Fig 2: The measured open-loop gain for Fig 1. Closed-loop gain is 27 dB, so feedback factor is easily calculated. 535b Fig 3: Generic amplifier THD plot, for 8 and 4 Ohm loading. Measurement bandwidth always 80 kHz, unless stated otherwise. Fig 4: Input stage transconductance against input voltage, for varying emitter degeneration resistances. Gain is lower but more constant for higher values. Fig 5: Input pair distortion from model amplifier. HF distortion is reduced as pair approaches balance, and is least when current - mirror enforces balance. 279b Fig 6: Test circuit for isolation of input distortion. Note the opamp works between 0V and -30V rails. Fig 7: Input distortion in isolation, showing that even a small Ic imbalance seriously increases distortion. Rise in curves below -10 dBu is due to noise floor. 223b Fig 8: Input stages, showing how value of R2 sets Ic balance. The third version with a degenerated current-mirror enforcing balance, gives the best results. Fig 9 The constant-gm degeneration technique. Both stages have the same transconductance, but the degenerated version is ten times more linear. Fig 10: VAS distortion in isolation, showing its reduction as the negative supply rail voltage is increased. Fig 11: VAS configurations. 11a, b show the two standard topologies. 11c, d are two methods for increasing local NFB through Cdom. 11e, f show VAS buffering. Fig 12: THD plot for a model amplifier at 15 Vrms. The middle trace shows an amplifier based on the small-signal section of Fig 33 the upper shows the extra VAS distortion without beta - enhancer TR12. The bottom trace is the distortion of the Audio Precision test system note the step at 20 kHz. 578b Fig 13: Standard Emitter-Follower, CFP, FET source-follower output stages. Fig 14: 14a: THD residual for underbiased Class-B, with spikes. 14b: Optimal-bias Class-B, showing discontinuity at crossover that cannot be removed. 14c: Class AB. Note the edges introduced by gm-doubling. Fig 15: Incremental gain diagram for Emitter-Follower output stages, loading from 16 to 2 Ohms Fig 16: Incremental gain diagram for CFP output stage, loading from 16 to 2 Ohms Fig 17: Incremental gain diagram for FET source-follower output stage, loading 16 to 2 Ohms. Fig 18 THD plot for a Blameless Class-B amplifier, 40W into 8 Ohms. Invar 1a. Fig 19 Large Signal Nonlinearity reduction by using sustained-beta output devices, doubled. 20W into 8 Ohms. Invar 23a. Fig 20 Crossover distortion with output stage underbiased by varying amounts. Lowest curve is for optimal biasing, and is essentially noise. 550 Fig 21 Crossover distortion at increasing frequencies for EF output stage. Note low-distortion area below -15 dB. (ref 25W8 Ohm) 551b Fig 22 Crossover distortion at increasing frequencies for CFP output stage. Low-distortion area is absent. 552b Fig 23 Crossover distortion under light loading. 68 Ohms is sufficient to produce measurable crossover. 541a Fig 24 The VAS collector impedance falls as frequency increases, due to local NFB through Cdom. Fig 25 VAS loading (Distortion 4) is present below 2 kHz if no measures to deal with it are taken. Fig 26 The varying input impedance of an EF output stage. Fig 27: Severe effect of misconnected rail decoupling: Distortion 5. Fig 28: Induction of half-wave currents: Distortion 6. Fig 29: Principle of NFB takeoff-point error: Distortion 7. Fig 30: The effect of NFB Takeoff-Point distortion. Fig 31: Electrolytic capacitor non-linearity for small sizes, eg in NFB arm: Distortion 8. Fig 32: Distortion from large electrolytic used as amplifier output capacitor: Distortion 8. 564a Fig 33: Circuit of Blameless power amplifier. Circuit changes from Fig 1 are minor. REFERENCES. 1 Lin, H C Transistor Audio Amplifier Electronics, Sept 1956, p173 2 Feucht Handbook of Analog Circuit Design Academic Press 1990, p256. (Pole-splitting) 3 Stocchino, G Letters, Electronics World, July 1995, p597. 4 Gray Meyer Analysis Design of Analog Integrated Circuits. Wiley 1984, p194. (tanh law of simple pair) 5 Gray Meyer Ibid, p256. (tanh law of current-mirror pair) 6 Gray Meyer Ibid, p251 (VAS law is portion of exponential) 7 Self, D Audio Power Amplifier Design Handbook. Newnes 1996, p231. ISBN 0-7506-2788-3 (poor FET linearity) 8 Self, D FETs vs BJTs - the linearity competition. Electronics Wireless World, May 1995 p387. (poor FET linearity) 9 Self, D Distortion In Power Amplifiers, Part 8. Electronics Wireless World, March 1994, p225 (Class A amp) 10 Self, D Thermal dynamics of Power Amplifiers: I Electronics World, May 1996, p410. 11 Self, D Thermal dynamics of Power Amplifiers: II Electronics World, June 1996, p481. 12 Self, D Thermal dynamics of Power Amplifiers: II Electronics World, Oct 1996, p754. 13 Self, D Load Invariant Audio Power Electronics World, Jan 1997, p16 (Beta droop) 14 Self, D Distortion In Power Amplifiers: Part 5. Electronics World, Dec 1993, p1011. (HF switchoff distortion losses) 15 Self, D Audio Power Amplifier Design Handbook. Newnes 1996, p128. ISBN 0-7506-2788-3 (VAS linearisation reduces collector impedance) 16 Self, D Distortion In Power Amplifiers: Part 3. Electronics World, Oct 1993, p820. (VAS linearisation) 17 Cherry, E A New Distortion Mechanism In Class-B amplifiers. JAES May 1981, p237. (Inductive distortion) 18 Baxandall, P Private communication, 1995 19 Self, D A Trimodal Power amplifier: I Wireless World, June 1995, p462 20 Self, D Load-Invariant Audio Power Electronics Wireless World, Jan 1997, p16 Douglas Self, London, Jan 1997. Words 9441The license information in the softkey, hardkey and VT hardware are all remotely upgradable. USB Hardkey for Multi-Instrument Series US39.95 Free Express Shipping It does not need a driver to run and thus is hassle free. It will be initialized to the license level purchased. With this option, you can run the software on the registered computer with the softkey, and on any computer with the hardkey. Full Package Pro all add-on modules, FREE upgrade for the same license level for life License Options: 1) Softkey activated license (locks to the registered computer) 2)VT hardware activated license (locks to the purchased VT hardware) Optionally, a USB hardkey (locks to the hardkey) can be purchased per softkey or VT hardware activated license 1. Introduction Multi-Instrument is a powerful multi-function virtual instrument software. It is a professional tool for time, frequency and time-frequency domain analysis. It supports a variety of hardware ranging from sound cards which are available in almost all computers to proprietary ADC and DAC hardware such as NI DAQmx cards, VT DSOs, VT RTAs and so on. It consists of the following instruments and functions. Oscilloscope Digital Oscilloscope Transient Recorder Data Recorder Voltmeter Lissajous Plot Digital Filters Persistence Mode Equivalent Time Sampling Signal Generator Function Generator Arbitrary Generator Burst Generator White Noise Generator Pink Noise Generator MultiTone Generator MLS Generator Musical Scale Generator DTMF Generator Frequency Sweep Amplitude Sweep Fade InFade Out DDS amp Streaming Modes DC Offset supported Multimeter Voltmeter, SPL Meter, Frequency Counter, RPM Meter, Counter, Duty Cycle Meter, FV Meter Spectrum Analyzer Amplitude Spectrum Analyzer Power Spectrum Analyzer Real Time Analyzer Octave Analyzer Phase Spectrum Analyzer Correlation Analyzer Freq. Response Measurement Distortion Analyzer Noise Analyzer Harmonics Analyzer Dynamic Signal Analyzer Coherence Measurement Transfer Function Measurement Impulse Response Measurement Spectrum 3D Plot Waterfall Plot Spectrogram Vibrometer Displacement, Velocity, Acceleration Conversion Data Logger 88 Derived Data Point Logger 151 Derived Data Points 16 User Defined Data Points Device Test Plan User Defined Plan 8 X-Y Plots 1 Test Report LCR Meter Inductor Meter Capacitor Meter Resistor Meter Impedance Meter DDP Viewer Display Derived Data Points HH, H, L, LL Alarming Software Customization amp Development Most Flexible Configuration ActiveX Automation Supported vtDAQ amp vtDAO Open Interfaces VC, VB, VC, Labview Samples System Requirement Windows XPVISTA7810, 32 or 64 bit, screen resolution 1024600 or higher. ADCDAC Hardware supported 81624 bit Windows compatible sound card (MMEASIO driver) NI DAQmx compatible cards VT DSO F1H1H2H3 series VT DAQ 12 series, VT DAO 1 series . Language supported English, French, German, Italian, Spanish, Portuguese, Russian Simplified Chinese, Traditional, Chinese, Japanese, Korean.
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